View AD8013_178012.PDF datasheet online --- IC-ON-LINE

Datasheet File OCR Text:

a
FEATURES Three Video Amplifiers in One Package Drives Large Capacitive Load Excellent Video Specifications (RL = 150 ) Gain Flatness 0.1 dB to 60 MHz 0.02% Differential Gain Error 0.06 Differential Phase Error Low Power Operates on Single +5 V to +13 V Power Supplies 4 mA/Amplifier Max Power Supply Current High Speed 140 MHz Unity Gain Bandwidth (3 dB) Fast Settling Time of 18 ns (0.1%) 1000 V/ s Slew Rate High Speed Disable Function per Channel Turn-Off Time 30 ns Easy to Use 95 mA Short Circuit Current Output Swing to Within 1 V of Rails APPLICATIONS LCD Displays Video Line Driver Broadcast and Professional Video Computer Video Plug-In Boards Consumer Video RGB Amplifier in Component Systems PRODUCT DESCRIPTION
Single Supply, Low Power, Triple Video Amplifier AD8013
PIN CONFIGURATION 14-Pin DIP & SOIC Package
DISABLE 1 DISABLE 2 DISABLE 3 +VS +IN 1 -IN 1 OUT 1 1 2 3 4 5 6 7 14 OUT 2 13 -IN 2 12 +IN 2
AD8013
11 -VS 10 +IN 3 9 8 -IN 3 OUT 3
differential gain and phase error of 0.02% and 0.06. This makes the AD8013 ideal for broadcast and professional video electronics. The AD8013 offers low power of 4 mA per amplifier max and runs on a single +5 V to +13 V power supply. The outputs of each amplifier swing to within one volt of either supply rail to easily accommodate video signals. The AD8013 is unique among current feedback op amps by virtue of its large capacitive load drive. Each op amp is capable of driving large capacitive loads while still achieving rapid settling time. For instance it can settle in 18 ns driving a resistive load, and achieves 40 ns (0.1%) settling while driving 200 pF. The outstanding bandwidth of 140 MHz along with 1000 V/s of slew rate make the AD8013 useful in many general purpose high speed applications where a single +5 V or dual power supplies up to 6.5 V are required. Furthermore the AD8013's high speed disable function can be used to power down the amplifier or to put the output in a high impedance state. This can then be used in video multiplexing applications. The AD8013 is available in the industrial temperature range of -40C to +85C.
500mV 500ns
The AD8013 is a low power, single supply, triple video amplifier. Each of the three amplifiers has 30 mA of output current, and is optimized for driving one back terminated video load (150 ) each. Each amplifier is a current feedback amplifier and features gain flatness of 0.1 dB to 60 MHz while offering
G = +2 RL = 150
0.2 0.1
NORMALIZED GAIN - dB
0 -0.1 -0.2 -0.3 -0.4 -0.5 1M VS = +5V
VS = 5V
100 9 0
1 0 0%
5V
100M 10M FREQUENCY - Hz 1G
Fine-Scale Gain Flatness vs. Frequency, G = +2, RL = 150
Channel Switching Characteristics for a 3:1 Mux
REV. A
Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. (c) Analog Devices, Inc., 1995 One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 617/329-4700 Fax: 617/326-8703
AD8013-SPECIFICATIONS (@ T = +25 C, R
A
LOAD
= 150
VS
, unless otherwise noted)
Min 100 110 AD8013A Typ Max 125 140 50 60 400 1000 18 40 Units MHz MHz MHz MHz V/s V/s ns ns
Model Conditions DYNAMIC PERFORMANCE Bandwidth (3 dB) Bandwidth (0.1 dB) Slew Rate Settling Time to 0.1% No Peaking, G = +2 No Peaking, G = +2 No Peaking, G = +2 No Peaking, G = +2 2 V Step 6 V Step 0 V to +2 V 4.5 V Step, CLOAD = 200 pF RLOAD > 1 k, RFB = 4 k fC = 5 MHz, RL = 1 k fC = 5 MHz, RL = 150 f = 10 kHz f = 10 kHz (-IIN) f = 3.58 MHz, G = +2 f = 3.58 MHz, G = +2
+5 V 5 V +5 V 5 V +5 V 5 V 5 V 6 V
600
NOISE/HARMONIC PERFORMANCE Total Harmonic Distortion Input Voltage Noise Input Current Noise Differential Gain (RL = 150 ) Differential Phase (RL = 150 ) DC PERFORMANCE Input Offset Voltage Offset Drift Input Bias Current (-) Input Bias Current (+) Open-Loop Transresistance
5 V 5 V +5 V, 5 V +5 V, 5 V +5 V1 5 V +5 V1 5 V +5 V, 5 V +5 V, 5 V +5 V, 5 V +5 V 5 V
-76 -66 3.5 12 0.05 0.02 0.06 0.06 2 7 2 3 800 1.1 M 650 200 150 2 3.8 1.2 52 52 56 56 0.2 5
0.05 0.12 5 10 15
dBc dBc nV/Hz pA/Hz % % Degrees Degrees mV V/C A A k k k k pF V +V dB dB A/V A/V
TMIN to TMAX TMIN to TMAX TMIN to TMAX TMIN to TMAX
650 550 800 k
INPUT CHARACTERISTICS Input Resistance Input Capacitance Input Common-Mode Voltage Range Common-Mode Rejection Ratio Input Offset Voltage Input Offset Voltage -Input Current +Input Current OUTPUT CHARACTERISTICS Output Voltage Swing RL = 1 k RL = 150 Output Current Short-Circuit Current Capacitive Load Drive MATCHING CHARACTERISTICS Dynamic Crosstalk Gain Flatness Match DC Input Offset Voltage -Input Bias Current
+Input -Input
5 V 5 V 5 V 5 V +5 V +5 V 5 V +5 V, 5 V +5 V, 5 V
3.8
0.4 7
VOL-VEE VCC-VOH VOL-VEE VCC-VOH +5 V 5 V 5 V 5 V 25
0.8 0.8 1.1 1.1 30 30 95 1000
1.0 1.0 1.3 1.3
V V V V mA mA mA pF
G = +2, f = 5 MHz f = 20 MHz
+5 V, 5 V 5 V +5 V, 5 V +5 V, 5 V
70 0.1 0.3 1.0
dB dB mV A
-2-
REV. A
AD8013
Model Conditions POWER SUPPLY Operating Range Quiescent Current/Amplifier Quiescent Current/Amplifier Power Supply Rejection Ratio Input Offset Voltage -Input Current +Input Current DISABLE CHARACTERISTICS Off Isolation Off Output Impedance Turn-On Time Turn-Off Time Switching Threshold Power Down VS = 2.5 V to 5 V Single Supply Dual Supply +5 V 5 V 6.5 V +5 V 5 V +5 V, 5 V +5 V, 5 V +5 V, 5 V +5 V, 5 V -VS + xV 1.3 70 VS Min +4.2 2.1 3.0 3.4 3.5 0.25 0.3 76 0.03 0.07 -70 12 50 30 1.6 AD8013A Typ Max +13 6.5 3.5 4.0 0.35 0.4 0.2 1.0 Units V V mA mA mA mA mA dB A/V A/V dB pF ns ns V
f = 6 MHz G = +1
1.9
NOTES 1 The test circuit for differential gain and phase measurements on a +5 V supply is ac coupled. Specifications subject to change without notice.
ABSOLUTE MAXIMUM RATINGS 1
Maximum Power Dissipation
Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . 13.2 V Total Internal Power Dissipation2 Plastic (N) . . . . . . . . . 1.6 Watts (Observe Derating Curves) Small Outline (R) . . . . 1.0 Watts (Observe Derating Curves) Input Voltage (Common Mode) . . Lower of VS or 12.25 V Differential Input Voltage . . . . . . . . Output 6 V (Clamped) Output Voltage Limit Maximum . . . . . . . . . Lower of (+12 V from -VS) or (+VS) Minimum . . . . . . . . . Higher of (-12.5 V from +VS) or (-VS) Output Short Circuit Duration . . . . . . . . . . . . . . . . . . . . Observe Power Derating Curves Storage Temperature Range N and R Package . . . . . . . . . . . . . . . . . . . -65C to +125C Operating Temperature Range AD8013A . . . . . . . . . . . . . . . . . . . . . . . . . . -40C to +85C Lead Temperature Range (Soldering 10 sec) . . . . . . . . +300C
NOTES 1 Stresses above those listed under "Absolute Maximum Ratings" may cause permanent damage to the device. This is a stress rating only and functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. 2 Specification is for device in free air: 14-Pin Plastic DIP Package: JA = 75C/Watt 14-Pin SOIC Package: JA = 120C/Watt
The maximum power that can be safely dissipated by the AD8013 is limited by the associated rise in junction temperature. The maximum safe junction temperature for the plastic encapsulated parts is determined by the glass transition temperature of the plastic, about 150C. Exceeding this limit temporarily may cause a shift in parametric performance due to a change in the stresses exerted on the die by the package. Exceeding a junction temperature of 175C for an extended period can result in device failure. While the AD8013 is internally short circuit protected, this may not be enough to guarantee that the maximum junction temperature is not exceeded under all conditions. To ensure proper operation, it is important to observe the derating curves. It must also be noted that in (noninverting) gain configurations (with low values of gain resistor), a high level of input overdrive can result in a large input error current, which may result in a significant power dissipation in the input stage. This power must be included when computing the junction temperature rise due to total internal power.
2.5 TJ = +150C
MAXIMUM POWER DISSIPATION - Watts
2.0 14-PIN DIP PACKAGE
ORDERING GUIDE
Model AD8013AN AD8013AR-14 AD8013AR-14-REEL AD8013AR-14-REEL7 AD8013ACHIPS Temperature Range -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C Package Description 14-Pin Plastic DIP 14-Pin Plastic SOIC 14-Pin Plastic SOIC 14-Pin Plastic SOIC Die Form Package Options N-14 R-14 R-14 R-14
1.5
14-PIN SOIC 1.0
0.5 -50 -40 -30 -20 -10 0 10 20 30 40 50 60 70 AMBIENT TEMPERATURE - C
80 90
Maximum Power Dissipation vs. Ambient Temperature
REV. A
-3-
AD8013
METALIZATION PHOTO
Contact factory for latest dimensions. Dimensions shown in inches and (mm).
+IN1 5 -IN1 6 +vs 4 DISABLE 3 3 2 DISABLE 2
OUT1 7
1 DISABLE 1
0.044 (1.13) 14 OUT 2 OUT3 8
-IN3 9 10 +IN3 0.071 (1.81) 11 -VS 12 +IN2 13 -IN2
CAUTION ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the human body and test equipment and can discharge without detection. Although the AD8013 features proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation or loss of functionality.
WARNING!
ESD SENSITIVE DEVICE
6
12
COMMON-MODE VOLTAGE RANGE - Volts
OUTPUT VOLTAGE SWING - V p-p
5
10 NO LOAD 8 RL = 150 6
4
3
2
4
1
2
0
1
2
3 4 5 SUPPLY VOLTAGE - Volts
6
7
0
1
2
3 4 5 SUPPLY VOLTAGE - Volts
6
7
Figure 1. Input Common-Mode Voltage Range vs. Supply Voltage
Figure 2. Output Voltage Swing vs. Supply Voltage
-4-
REV. A
AD8013
10 VS = 5V
INPUT BIAS CURRENT - A
3
OUTPUT VOLTAGE SWING - V p-p
8
2
1
6
0 -IB -1
4 VS = +5V 2
-2 +IB
0 10
1k 100 LOAD RESISTANCE -
10k
-3 -60
-40
-20
0 20 40 60 80 100 JUNCTION TEMPERATURE - C
120
140
Figure 3. Output Voltage Swing vs. Load Resistance
Figure 6. Input Bias Current vs. Junction Temperature
12
2
SUPPLY CURRENT - mA
VS = 5V 10
INPUT OFFSET VOLTAGE - mV
11
1
0
9 VS = +5V 8
-1 VS = +5V -2 VS = 5V -3
7
6 -60
-40
-20
0 20 40 60 80 100 JUNCTION TEMPERATURE - C
120
140
-4 -60
-40 -20
0 20 40 60 80 100 JUNCTION TEMPERATURE - C
120
140
Figure 4. Total Supply Current vs. Junction Temperature
Figure 7. Input Offset Voltage vs. Junction Temperature
11
140 VS = 5V TA = +25C
SHORT CIRCUIT CURRENT - mA
130 SOURCE 120 SINK 100
SUPPLY CURRENT - mA
10
9
8
90
7
1
2
3 4 5 SUPPLY VOLTAGE - Volts
6
7
80 -60
-40
-20
0 20 40 60 80 100 JUNCTION TEMPERATURE - C
120
140
Figure 5. Supply Current vs. Supply Voltage
Figure 8. Short Circuit Current vs. Junction Temperature
REV. A
-5-
AD8013
1k
CLOSED-LOOP OUTPUT RESISTANCE -
70
G = +2 100
COMMON-MODE REJECTION - dB
60 VCM
R R R
R
50
10
40
1
VS = 5V
30
0.1
20
0.01 100k
1M
10M FREQUENCY - Hz
100M
1G
10 100k
1M
10M FREQUENCY - Hz
100M
1G
Figure 9. Closed-Loop Output Resistance vs. Frequency
Figure 12. Common-Mode Rejection vs. Frequency
100k
80 70 VS = 5V
10k
POWER SUPPLY REJECTION - dB
OUTPUT RESISTANCE -
60 50 +PSR 40 30 20 10 -PSR
1k
100
10 1M
100M 10M FREQUENCY - Hz
1G
0 100k
1M
10M FREQUENCY - Hz
100M
1G
Figure 10. Output Resistance vs. Frequency, Disabled State
Figure 13. Power Supply Rejection Ratio vs. Frequency
1k
1k
VS = 5V RL = 1k
-45 -90 -135
VOLTAGE NOISE nV/Hz
CURRENT NOISE pA/Hz
120
100
100
TRANSIMPEDANCE - dB
100
-180
NONINVERTING I 10 INVERTING I 10
80
60
VNOISE 1 100 1 1k 10k FREQUENCY - Hz 100k 1M
40 10k
100k
1M 10M FREQUENCY - Hz
100M
1G
Figure 11. Input Current and Voltage Noise vs. Frequency
Figure 14. Open-Loop Transimpedance vs. Frequency (Relative to 1 )
-6-
REV. A
PHASE - Degrees
140
0
AD8013
-40
HARMONIC DISTORTION - dBc
-50 -60 -70 -80 -90 -100 -110 -120 1k
G = +2 VO = 2V p-p VS = 5V +1 0 2nd RL = 150
PHASE VS = +5V
G = +1 RL = 150 VS = 5V
0 -90 -180 -270
CLOSED-LOOP GAIN (NORMALIZED) - dB
GAIN VS = 5V
-1 -2 -3 VS = +5V -4 -5
2nd RL = 1k 3rd RL = 150 10k 100k 1M FREQUENCY - Hz
3rd RL = 1k
10M
100M
-6 1M
100M 10M FREQUENCY - Hz
1G
Figure 15. Harmonic Distortion vs. Frequency
Figure 18. Closed-Loop Gain and Phase vs. Frequency, G = +1, RL = 150
1800 1600 1400 VS = 5V RL = 500
2000 1800
G = +10
SLEW RATE - V/s
G = +10 1600 1400 G = -1 1200 G = +2 1000 800 G = +1 600 400 200
SLEW RATE - V/s
1200 G = -1 1000 800 600 400 200 1 2 3 4 5 6 OUTPUT STEP SIZE - V p-p 7 8 G = +2
G = +1
1.5
2.5
3.5 4.5 5.5 SUPPLY VOLTAGE - Volts
6.5
7.5
Figure 16. Slew Rate vs. Output Step Size
Figure 19. Maximum Slew Rate vs. Supply Voltage
2V
100
20ns
100
500mV
VIN
90
20ns
VIN
90
VOUT
10 0%
VOUT
10 0%
2V
500mV
Figure 17. Large Signal Pulse Response, Gain = +1, (RF = 2 k, RL = 150 , VS = 5 V)
Figure 20. Small Signal Pulse Response, Gain = +1, (RF = 2 k, RL = 150 , VS = 5 V)
REV. A
-7-
PHASE SHIFT - Degrees
-30
AD8013
50mV
100
20ns
100
2V
VIN
90
20ns
VIN
90
VOUT
10 0%
VOUT
10 0%
500mV
2V
Figure 21. Large Signal Pulse Response, Gain = +10, RF = 301 , RL = 150 , VS = 5 V)
Figure 24. Large Signal Pulse Response, Gain = -1, (RF = 698 , RL = 150 , VS = 5 V)
PHASE SHIFT - Degrees
PHASE VS = +5V +1
G = +10 RL = 150 VS = 5V
0 -90 -180
PHASE VS = +5V
G = -1 RL = 150 VS = 5V
180 90 0 -90
+1
CLOSED-LOOP GAIN (NORMALIZED) - dB
0 -1 -2 -3 -4 -5 -6 1M
CLOSED-LOOP GAIN (NORMALIZED) - dB
GAIN
-270
0 -1 -2 -3 -4 -5
GAIN
VS = +5V VS = 5V
VS = 5V VS = +5V
10M 100M FREQUENCY - Hz
1G
-6 1M
10M 100M FREQUENCY - Hz
1G
Figure 22. Closed-Loop Gain and Phase vs. Frequency, G = +10, RL = 150
Figure 25. Closed-Loop Gain and Phase vs. Frequency, G = -1, RL = 150
50mV
100
20ns
100
500mV
VIN
90
20ns
VIN
90
VOUT 10
0%
VOUT
10 0%
500mV
500mV
Figure 23. Small Signal Pulse Response, Gain = +10, (RF = 301 , RL = 150 , VS = 5 V)
Figure 26. Small Signal Pulse Response, Gain = -1, (RF = 698 , RL = 150 , VS = 5 V)
-8-
REV. A
PHASE SHIFT - Degrees
AD8013
G = -10 RL = 150 VS = 5V +1 VS = +5V 180 90 0 -90
PHASE
CLOSED-LOOP GAIN (NORMALIZED) - dB
0 -1 -2 -3
GAIN
VS = +5V -4 -5 -6 1M 100M 10M FREQUENCY - Hz
VS = 5V
To estimate the -3 dB bandwidth for closed-loop gains of 2 or greater, for feedback resistors not listed in the following table, the following single pole model for the AD8013 may be used: G ACL 1 + SC (R + Gn rin ) T F where: CT = transcapacitance 1 pF RF = feedback resistor G = ideal closed loop gain RF Gn = 1 + R = noise gain G rin = inverting input resistance 150 ACL = closed loop gain The -3 dB bandwidth is determined from this model as: 1 f3 2 CT (RF + Gn rin ) This model will predict -3 dB bandwidth to within about 10% to 15% of the correct value when the load is 150 and VS = 5 V. For lower supply voltages there will be a slight decrease in bandwidth. The model is not accurate enough to predict either the phase behavior or the frequency response peaking of the AD8013. It should be noted that the bandwidth is affected by attenuation due to the finite input resistance. Also, the open-loop output resistance of about 12 reduces the bandwidth somewhat when driving load resistors less than about 250 . (Bandwidths will be about 10% greater for load resistances above a few hundred ohms.)
Table I. -3 dB Bandwidth vs. Closed-Loop Gain and Feedback Resistor, RL = 150 (SOIC) VS - Volts 5 Gain +1 +2 +10 -1 -10 +1 +2 +10 -1 -10 RF - Ohms 2000 845 (931) 301 698 (825) 499 2000 887 (931) 301 698 (825) 499 BW - MHz 230 150 (135) 80 140 (130) 85 180 120 (130) 75 130 (120) 80
1G
Figure 27. Closed-Loop Gain and Phase vs. Frequency, G = -10, RL = 150
General
The AD8013 is a wide bandwidth, triple video amplifier that offers a high level of performance on less than 4.0 mA per amplifier of quiescent supply current. The AD8013 uses a proprietary enhancement of a conventional current feedback architecture, and achieves bandwidth in excess of 200 MHz with low differential gain and phase errors, making it an extremely efficient video amplifier. The AD8013's wide phase margin coupled with a high output short circuit current make it an excellent choice when driving any capacitive load. High open-loop gain and low inverting input bias current enable it to be used with large values of feedback resistor with very low closed-loop gain errors. It is designed to offer outstanding functionality and performance at closed-loop inverting or noninverting gains of one or greater.
Choice of Feedback & Gain Resistors
Because it is a current feedback amplifier, the closed-loop bandwidth of the AD8013 may be customized using different values of the feedback resistor. Table I shows typical bandwidths at different supply voltages for some useful closed-loop gains when driving a load of 150 . The choice of feedback resistor is not critical unless it is important to maintain the widest, flattest frequency response. The resistors recommended in the table are those (chip resistors) that will result in the widest 0.1 dB bandwidth without peaking. In applications requiring the best control of bandwidth, 1% resistors are adequate. Package parasitics vary between the 14-pin plastic DIP and the 14-pin plastic SOIC, and may result in a slight difference in the value of the feedback resistor used to achieve the optimum dynamic performance. Resistor values and widest bandwidth figures are shown in parenthesis for the SOIC where they differ from those of the DIP. Wider bandwidths than those in the table can be attained by reducing the magnitude of the feedback resistor (at the expense of increased peaking), while peaking can be reduced by increasing the magnitude of the feedback resistor. Increasing the feedback resistor is especially useful when driving large capacitive loads as it will increase the phase margin of the closed-loop circuit. (Refer to the section on driving capacitive loads for more information.) REV. A -9-
PHASE SHIFT - Degrees
+5
Driving Capacitive Loads
When used in combination with the appropriate feedback resistor, the AD8013 will drive any load capacitance without oscillation. The general rule for current feedback amplifiers is that the higher the load capacitance, the higher the feedback resistor required for stable operation. Due to the high open-loop transresistance and low inverting input current of the AD8013, the use of a large feedback resistor does not result in large closedloop gain errors. Additionally, its high output short circuit current makes possible rapid voltage slewing on large load capacitors. For the best combination of wide bandwidth and clean pulse response, a small output series resistor is also recommended. Table II contains values of feedback and series resistors which result in the best pulse responses. Figure 29 shows the AD8013 driving a 300 pF capacitor through a large voltage step with virtually no overshoot. (In this case, the large and small signal pulse responses are quite similar in appearance.)
AD8013
RF +VS RG
4
1.0F 0.1F 15 1.0F 0.1F RS VO CL
AD8013
VIN RT -VS
11
As noted in the warning under "Maximum Power Dissipation," a high level of input overdrive in a high noninverting gain circuit can result in a large current flow in the input stage. Though this current is internally limited to about 30 mA, its effect on the total power dissipation may be significant.
High Performance Video Line Driver
Figure 28. Circuit for Driving a Capacitive Load
Table II. Recommended Feedback and Series Resistors vs. Capacitive Load and Gain CL - pF 20 50 100 200 300 500 RF - Ohms 2k 2k 3k 4k 6k 7k
500mV
100
At a gain of +2, the AD8013 makes an excellent driver for a back terminated 75 video line (Figures 31, 32, and 33). Low differential gain and phase errors and wide 0.1 dB bandwidth can be realized. The low gain and group delay matching errors ensure excellent performance in RGB systems. Figures 34 and 35 show the worst case matching.
RG RF +VS
4
RS - Ohms G=2 G3 25 25 20 15 15 15 15 15 15 15 15 15
75 CABLE VIN
0.1F 75 75 CABLE VOUT 0.1F 75
AD8013
11
75 -VS
50ns
Figure 31. A Video Line Driver Operating at a Gain of +2 (RF = RG from Table I)
G = +2 RL = 150 VS = 5V +1 VS = +5V 0 -90 -180 -270 VS = 5V
VIN
90
PHASE
0%
CLOSED-LOOP GAIN (NORMALIZED) - dB
VOUT
10
0 -1 -2 -3
GAIN
1V
Figure 29. Pulse Response Driving a Large Load Capacitor. CL = 300 pF, G = +2, RF = 6k, RS = 15
Overload Recovery
VS = +5V -4 -5 -6 1M 100M 10M FREQUENCY - Hz 1G
NORMALIZED GAIN - dB
The three important overload conditions are: input commonmode voltage overdrive, output voltage overdrive, and input current overdrive. When configured for a low closed-loop gain, the amplifier will quickly recover from an input commonmode voltage overdrive; typically in under 25 ns. When configured for a higher gain, and overloaded at the output, the recovery time will also be short. For example, in a gain of +10, with 15% overdrive, the recovery time of the AD8013 is about 20 ns (see Figure 30). For higher overdrive, the response is somewhat slower. For 6 dB overdrive, (in a gain of +10), the recovery time is about 65 ns.
500mV
100
Figure 32. Closed-Loop Gain & Phase vs. Frequency for the Line Driver
G = +2 RL = 150 +0.2 +0.1 0 -0.1 -0.2 -0.3 -0.4 -0.5 1M 10M 100M FREQUENCY - Hz 1G VS = +5V VS = 5V
50ns
VIN
90
VOUT
10 0%
5V
Figure 33. Fine-Scale Gain Flatness vs. Frequency, G = +2, RL = 150
Figure 30. 15% Overload Recovery, G = +10 (RF = 300 , RL = 1 k, VS = 5 V)
-10-
REV. A
PHASE SHIFT - Degrees
AD8013
1.5 1.0 0.5 G = +2 RL = 150
VI 8k
+5V
GAIN MATCHING - dB
TO DISABLE PIN 4k 10k -5V
VS = +5V
0 -0.5 VS = 5V -1.0
V I HIGH => AMPLIFIER ENABLED V I LOW => AMPLIFIER DISABLED
Figure 36. Level Shifting to Drive Disable Pins on Dual Supplies
-1.5 -2.0 1M
100M 10M FREQUENCY - Hz
1G
The AD8013's input stages include protection from the large differential input voltages that may be applied when disabled. Internal clamps limit this voltage to about 3 V. The high input to output isolation will be maintained for voltages below this limit.
3:1 Video Multiplexer
Figure 34. Closed-Loop Gain Matching vs. Frequency
10 8 6 GROUP DELAY - ns 4 2 1.0 0.5 0 -0.5 -1.0 100k DELAY MATCHING 10M 1M FREQUENCY - Hz VS = +5V G = +2 RL = 150 VS = 5V VS = 5V VS = +5V G = +2 RL = 150
Wiring the amplifier outputs together will form a 3:1 mux with excellent switching behavior. Figure 37 shows a recommended configuration which results in -0.1 dB bandwidth of 35 MHz and OFF channel isolation of 60 dB at 10 MHz on 5 V supplies. The time to switch between channels is about 50 ns. Switching time is virtually unaffected by signal level.
DELAY
665 845 +VS
6 4 7
84
VIN1
5
1
100M
DISABLE 1
75
Figure 35. Group Delay and Group Delay Matching vs. Frequency, G = +2, RL = 150
Disable Mode Operation
VIN2
665
845 75 CABLE VOUT 75
13 14 12 2
84
Pulling the voltage on any one of the Disable pins about 1.6 V up from the negative supply will put the corresponding amplifier into a disabled, powered down, state. In this condition, the amplifier's quiescent current drops to about 0.3 mA, its output becomes a high impedance, and there is a high level of isolation from input to output. In the case of the gain of two line driver for example, the impedance at the output node will be about the same as for a 1.6 k resistor (the feedback plus gain resistors) in parallel with a 12 pF capacitor and the input to output isolation will be about 66 dB at 5 MHz. Leaving the Disable pin disconnected (floating) will leave the corresponding amplifier operational, in the enabled state. The input impedance of the disable pin is about 40 k in parallel with a few picofarads. When driven to 0 V, with the negative supply at -5 V, about 100 A flows into the disable pin. When the disable pins are driven by complementary output CMOS logic, on a single 5 V supply, the disable and enable times are about 50 ns. When operated on dual supplies, level shifting will be required from standard logic outputs to the Disable pins. Figure 36 shows one possible method which results in a negligible increase in switching time.
75 DISABLE 2 665 845
9 8 11
84
3
VIN3 75
10
-VS DISABLE 3
Figure 37. A Fast Switching 3:1 Video Mux (Supply Bypassing Not Shown)
500mV
100 90
200ns
10 0%
5V
Figure 38. Channel Switching Characteristic for the 3:1 Mux
REV. A
-11-
AD8013
2:1 Video Multiplexer
698
698 +5V 6 5
13
R1 2k
2k 13
4 1 7
2
VINA
12 2
14
R3 10 DISABLE
5
100 VIN 12 50
14 DIS 1 1k 10 1k 9 DIS 3 3 11 -5V 845 845 8 15 VOUT
VINB
10
R4 10
1
6
7
VOUT
3
9 3
8
R2 2k DISABLE
R5 845
R6 845
Figure 39. 2:1 Mux with High Isolation and Low Differential Gain and Phase Errors
2 1 GAIN 0
CLOSED-LOOP GAIN - dB
100 90
Figure 41. Circuit to Switch Between Gains of -1 and +1
500mV 500mV 200ns
-1 -2 -3 -4 -5 FEEDTHROUGH -6 -7 -8 1M 100M 10M FREQUENCY - Hz -60 -70 -80 1G -30 -40 -50
FEEDTHROUGH - dB
10 0%
5V
Figure 42. Switching Characteristic for Circuit of Figure 41
Figure 40. 2:1 Mux ON Channel Gain and Mux OFF Channel Feedthrough vs. Frequency
Gain Switching
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).
14-Lead Plastic DIP (N-14)
0.795 (20.19) 0.725 (18.42)
14 1 8 7
14-Lead SOIC (R-14)
0.3444 (8.75) 0.3367 (8.55)
0.280 (7.11) 0.240 (6.10) 0.325 (8.25) 0.300 (7.62) 0.195 (4.95) 0.115 (2.93)
0.1574 (4.00) 0.1497 (3.80)
14 1
8 7
0.2440 (6.20) 0.2284 (5.80)
PIN 1 0.210 (5.33) MAX 0.160 (4.06) 0.115 (2.93) 0.022 (0.558) 0.014 (0.356)
0.060 (1.52) 0.015 (0.38) 0.130 (3.30) MIN 0.100 0.070 (1.77) (2.54) 0.045 (1.15) BSC SEATING PLANE
PIN 1 0.0098 (0.25) 0.0040 (0.10)
0.0688 (1.75) 0.0532 (1.35)
0.0196 (0.50) x 45 0.0099 (0.25)
0.015 (0.381) 0.008 (0.204)
SEATING PLANE
0.0500 (1.27) BSC
0.0192 (0.49) 0.0138 (0.35)
0.0098 (0.25) 0.0075 (0.19)
8 0
0.0500 (1.27) 0.0160 (0.41)
-12-
REV. A
PRINTED IN U.S.A.
C2084-18-10/95
Configuring two amplifiers as unity gain followers and using the third to set the gain results in a high performance 2:1 mux (Figures 39 and 40). This circuit takes advantage of the very low crosstalk between Channels 2 and 3 to achieve the OFF channel isolation shown in Figure 40. This circuit can achieve differential gain and phase of 0.03% and 0.07 respectively.
The AD8013 can be used to build a circuit for switching between any two arbitrary gains while maintaining a constant input impedance. The example of Figure 41 shows a circuit for switching between a noninverting gain of 1 and an inverting gain of 1. The total time for channel switching and output voltage settling is about 80 ns.

▲Up To Search▲

Price & Availability of AD8013

	To Download AD8013 Datasheet File
If you can't view the Datasheet, Please click here to try to view without PDF Reader .